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19-2091; Rev 0; 8/01
High-Efficiency, Current-Mode, Inverting PWM Controller
General Description
MAX1846/MAX1847 high-efficiency PWM inverting controllers allow designers to implement compact, lownoise, negative-output DC-DC converters for telecom and networking applications. Both devices operate from +3V to +16.5V input and generate -2V to -200V output. To minimize switching noise, both devices feature a current-mode, constant-frequency PWM control scheme. The operating frequency can be set from 100kHz to 500kHz through a resistor. The MAX1846 is available in an ultra-compact 10-pin MAX package. Operation at high frequency, compatibility with ceramic capacitors, and inverting topology without transformers allow for a compact design. Compatibility with electrolytic capacitors and flexibility to operate down to 100kHz allow users to minimize the cost of external components. The high-current output drivers are designed to drive a P-channel MOSFET and allow the converter to deliver up to 30W. The MAX1847 features clock synchronization and shutdown functions. The MAX1847 can also be configured to operate as an inverting flyback controller with an Nchannel MOSFET and a transformer to deliver up to 70W. The MAX1847 is available in a 16-pin QSOP package. Current-mode control simplifies compensation and provides good transient response. Accurate current-mode control and over current protection are achieved through low-side current sensing. o 90% Efficiency o +3.0V to +16.5V Input Range o -2V to -200V Output o Drives High-Side P-Channel MOSFET o 100kHz to 500kHz Switching Frequency o Current-Mode, PWM Control o Internal Soft-Start o Electrolytic or Ceramic Output Capacitor o The MAX1847 also offers: Synchronization to External Clock Shutdown N-Channel Inverting Flyback Option
Features
MAX1846/MAX1847
Ordering Information
PART MAX1846EUB MAX1847EEE TEMP. RANGE -40C to +85C -40C to +85C PIN-PACKAGE 10 MAX 16 QSOP
Typical Operating Circuit
POSITIVE VIN
Applications
Cellular Base Stations Networking Equipment Optical Networking Equipment SLIC Supplies CO DSL Line Driver Supplies Industrial Power Supplies Automotive Electronic Power Supplies Servers
FREQ PGND VL IN EXT P NEGATIVE VOUT
MAX1846 MAX1847
COMP CS
REF GND FB
Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
ABSOLUTE MAXIMUM RATINGS
IN, SHDN to GND ...................................................-0.3V to +20V PGND to GND .......................................................-0.3V to +0.3V VL to PGND for VIN 5.7V...........................-0.3V to (VIN + 0.3V) VL to PGND for VIN > 5.7V .......................................-0.3V to +6V EXT to PGND ...............................................-0.3V to (VIN + 0.3V) REF, COMP to GND......................................-0.3V to (VL + 0.3V) CS, FB, FREQ, POL, SYNC to GND .........................-0.3V to +6V Continuous Power Dissipation (TA = +70C) 10-Pin MAX (derate 5.6mW/C above +70C) ...........444mW 16-Pin QSOP (derate 8.3mW/C above +70C)...........696mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147k 1%, CVL = 0.47F, CREF = 0.1F, TA = 0C to +85C, unless otherwise noted.)
PARAMETER PWM CONTROLLER Operating Input Voltage Range UVLO Threshold UVLO Hysteresis FB Threshold FB Input Current Load Regulation Line Regulation Current-Limit Threshold CS Input Current Supply Current Shutdown Supply Current REF Output Voltage REF Load Regulation VL Output Voltage VL Load Regulation CS = GND VFB = -0.1V, VIN = +3.0V to +16.5V SHDN = GND, VIN = +3.0V to +16.5V IREF = 50A IREF = 0 to 500A IVL = 100A IVL = 0.1mA to 2.0mA 3.85 1.236 No load VFB = -0.1V CCOMP = 0.068F, VOUT = -48V, IOUT = 20mA to 200mA (Note 1) CCOMP = 0.068F, VOUT = -48V, VIN = +8V to +16.5V, IOUT = 100mA 85 -12 -50 -1 0.04 100 10 0.75 10 1.25 -2 4.25 -20 115 20 1.2 25 1.264 -15 4.65 -60 VIN rising VIN falling 2.6 3.0 2.8 2.74 60 0 -6 12 50 0 16.5 2.95 V V mV mV nA % % mV A mA A V mV V mV CONDITIONS MIN TYP MAX UNITS
REFERENCE AND VL REGULATOR
2
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High-Efficiency, Current-Mode, Inverting PWM Controller
ELECTRICAL CHARACTERISTICS (continued)
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147k 1%, CVL = 0.47F, CREF = 0.1F, TA = 0C to +85C, unless otherwise noted.)
OSCILLATOR RFREQ = 500k 1% Oscillator Frequency RFREQ = 147k 1% RFREQ = 76.8k 1% RFREQ = 500k 1% Maximum Duty Cycle SYNC Input Signal Duty-Cycle Range Minimum SYNC Input Logic Low Pulse Width SYNC Input Rise/Fall Time SYNC Input Frequency Range DIGITAL INPUTS POL, SYNC, SHDN Input High Voltage POL, SYNC, SHDN Input Low Voltage POL, SYNC Input Current SHDN Input Current SOFT-START Soft-Start Clock Cycles Soft-Start Levels EXT OUTPUT EXT Sink/Source Current EXT On-Resistance VIN = +5V, VEXT forced to +2.5V EXT high or low, tested with 100mA load, VIN = +5V EXT high or low, tested with 100mA load, VIN = +3V 1 2 5 5 10 A 1024 64 Cycles POL, SYNC = GND or VL VSHDN = +5V or GND VSHDN = +16.5V -12 20 -4 1.5 2.0 0.45 40 0 6 V V A A (Note 2) 100 RFREQ = 147k 1% RFREQ = 76.8kz 1% 7 50 93 85 90 255 100 300 500 96 88 80 93 200 200 550 % ns ns kHz 97 90 % 110 345 kHz
MAX1846/MAX1847
Note 1: Production test correlates to operating conditions. Note 2: Guaranteed by design and characterization.
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
ELECTRICAL CHARACTERISTICS
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147k 1%, CVL = 0.47F, CREF = 0.1F, TA = -40C to +85C, unless otherwise noted.) (Note 3)
PARAMETER PWM CONTROLLER Operating Input Voltage Range UVLO Threshold FB Threshold FB Input Current Load Regulation Current Limit Threshold CS Input Current Supply Current Shutdown Supply Current REF Output Voltage REF Load Regulation VL Output Voltage VL Load Regulation OSCILLATOR Oscillator Frequency Maximum Duty Cycle SYNC Input Signal Duty-Cycle Range Minimum SYNC Input Logic Low Pulse Width SYNC Input Rise/Fall Time SYNC Input Frequency Range DIGITAL INPUTS POL, SYNC, SHDN Input High Voltage POL, SYNC, SHDN Input Low Voltage 2.0 0.45 V V (Note 2) 100 RFREQ = 500k 1% RFREQ = 147k 1% RFREQ = 500k 1% RFREQ = 147k 1% 84 255 93 84 7 116 345 98 93 93 200 200 550 kHz % % ns ns kHz CS = GND VFB = -0.1V, VIN = +3.0V to +16.5V SHDN = GND, VIN = +3.0V to +16.5V IREF = 50A IREF = 0 to 500A IVL = 100A IVL = 0.1mA to 2.0mA 3.85 1.225 VIN rising VIN falling No load VFB = -0.1V CCOMP = 0.068F, VOUT = -48V, IOUT= 20mA to 200mA (Note 1) 2.6 -20 -50 -2 85 20 50 0 115 20 1.2 25 1.275 -15 4.65 -60 3.0 16.5 2.95 V V mV nA % mV A mA A V mV V mV CONDITIONS MIN MAX UNITS
REFERENCE AND VL REGULATOR
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
ELECTRICAL CHARACTERISTICS (continued)
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147k 1%, CVL = 0.47F, CREF = 0.1F, TA = -40C to +85C, unless otherwise noted.) (Note 3)
PARAMETER POL, SYNC Input Current SHDN Input Current EXT OUTPUT EXT On-Resistance EXT high or low, IEXT = 100mA, VIN = +5V EXT high or low, IEXT = 100mA, VIN = +3V 7.5 12 CONDITIONS POL, SYNC = GND or VL V SHDN = +5V or GND V SHDN = +16.5V -12 MIN MAX 40 0 6 UNITS A A
Note 3: Parameters to -40C are guaranteed by design and characterization.
Typical Operating Characteristics
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47F, CREF = 0.1F, TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT
MAX1846/7 toc01
EFFICIENCY vs. LOAD CURRENT
VIN = 5V
MAX1846/7 toc02
EFFICIENCY vs. LOAD CURRENT
90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 VIN = 16.5V VIN = 12V
MAX1846/7 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 1 APPLICATION CIRCUIT A 10 100 VOUT = -5V 1000 VIN = 16.5V VIN = 5V
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 APPLICATION CIRCUIT B 1 10 100 VIN = 3V
100
VIN = 3.3V
VOUT = -12V 1000 10,000
0 1
APPLICATION CIRCUIT C 10 100
VOUT = -48V 1000
10,000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
LOAD CURRENT (mA)
OUTPUT VOLTAGE LOAD REGULATION
MAX1846/7 toc04
SUPPLY CURRENT vs. SUPPLY VOLTAGE
1.4 1.2 1.0 IIN (mA) 0.8 0.6 0.4 0.2 1.242 VFB = -0.1V 0 1.238 16 -40 -20
MAX1846/7 toc05
REFERENCE VOLTAGE vs. TEMPERATURE
MAX1846/7 toc06
-11.90 -11.92 -11.94 OUTPUT VOLTAGE (V) -11.96 -11.98 -12.00 -12.02 -12.04 -12.06 -12.08 -12.10 0 APPLICATION CIRCUIT B 100 200 300 VIN = 5V 400 500
1.6
1.262 1.258 1.254 VREF (V) 1.250 1.246
600
0
2
4
6
8 VIN (V)
10
12
14
0
20
40
60
80
100
LOAD CURRENT (mA)
TEMPERATURE (C)
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47F, CREF = 0.1F, TA = +25C, unless otherwise noted.)
VL VOLTAGE vs. TEMPERATURE
MAX1846/7 toc07
REFERENCE LOAD REGULATION
1.260 4.340 4.300 4.260 VREF (V) 1.250 VL (V) 4.220 4.180 1.245 4.140
VL LOAD REGULATION
MAX1846/7 toc08 MAX1846/7 toc09
4.27
4.26
1.255
VL (V) IVL = 0
4.25
4.24
4.23
1.240 0 100 200 300 400 500 IREF (A)
4.100 -40 -20 0 20 40 60 80 100 TEMPERATURE (C)
4.22 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 IVL (mA)
SHUTDOWN SUPPLY CURRENT vs. TEMPERATURE
MAX1846/7 toc10
OPERATING CURRENT vs. TEMPERATURE
MAX1846/7 toc11
SWITCHING FREQUENCY vs. RFREQ
A
MAX1846/7 toc12
16 SHUTDOWN SUPPLY CURRENT (A) 14 12 10 8 6 4 2 0 -40 -20 0 60 40 TEMPERATURE (C) 20 80 VIN = 3V VIN = 16.5V VIN = 10V
14 A: VIN = 3V, VOUT = -12V 12 OPERATING CURRENT (mA) 10 8 6 4 APPLICATION CIRCUIT A B: VIN = 5V, VOUT = -5V C: VIN = 16.5V, VOUT = -5V
500
400
fOSC (kHz) B C 60 80 100
300
200
100 2 0 100 -40 -20 0 20 40 TEMPERATURE (C)
0 0 100 200 300 RFREQ (k) 400 500 600
SWITCHING FREQUENCY vs. TEMPERATURE
MAX1846/7 toc13
EXT RISE/FALL TIME vs. CAPACITANCE
MAX1846/7 toc14
EXITING SHUTDOWN
MAX1846/7 toc15
302 301 300 FREQUENCY (kHz) 299 298 297 296 295 294 -40 -20 0 20 RFREQ = 147k 1% 60 40 TEMPERATURE (C) 80
160 140 120 100 TIME (ns) 80 60 40 RISE TIME 20 0 VIN = 12V 0 2000 4000 6000 8000 FALL TIME
5V/div SHDN 0 VOUT 5V/div
IL APPLICATION CIRCUIT B 1ms/div
1A/div
100
10,000
CAPACITANCE (pF)
6
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High-Efficiency, Current-Mode, Inverting PWM Controller
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47F, CREF = 0.1F, TA = +25C, unless otherwise noted.)
ENTERING SHUTDOWN
MAX1846/7 toc16
MAX1846/MAX1847
HEAVY-LOAD SWITCHING WAVEFORM
MAX1846/7 toc17
SHDN 5V/div VOUT 100mV/div
0
5V/div VOUT IL
1A/div
IL APPLICATION CIRCUIT B 1ms/div
1A/div
LX
10V/div
APPLICATION CIRCUIT B 1s/div ILOAD = 600mA
LIGHT-LOAD SWITCHING WAVEFORM
MAX1846/7 toc18
VOUT
100mV/div
1A/div IL
LX
10V/div
APPLICATION CIRCUIT B 1s/div ILOAD = 50mA
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc19
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc20
ILOAD
ILOAD
VOUT 500mV/div
VOUT
200mV/div
IL
1A/div
IL
500mA/div
APPLICATION CIRCUIT B 2ms/div ILOAD = 10mA to 400mA
APPLICATION CIRCUIT C 400s/div ILOAD = 4mA to 100mA
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
Pin Description
PIN MAX1846 -- 1 MAX1847 1 2 NAME FUNCTION Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS lowside FET in transformer-based applications. VL Low-Dropout Regulator. Connect 0.47F ceramic capacitor from VL to GND. Oscillator Frequency Set Input. Connect a resistor (RFREQ) from FREQ to GND to set the internal oscillator frequency from 100kHz (RFREQ = 500k) to 500kHz (RFREQ = 76.8k). RFREQ is still required if an external clock is used at SYNC. See Setting the Operating Frequency section. Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network from COMP to GND for loop compensation. See Design Procedure. 1.25V Reference Output. REF can source up to 500A. Bypass with a 0.1F ceramic capacitor from REF to GND. Feedback Input. Connect FB to the center of a resistor-divider connected between the output and REF. The FB threshold is 0. No Connection Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect to IN for normal operation. Analog Ground. Connect to PGND. Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND. Positive Current-Sense Input. Connect a current-sense resistor (RCS) between CS and External MOSFET Gate-Driver Output. EXT swings from IN to PGND. Power-Supply Input Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set the internal oscillator frequency with RFREQ. Drive SYNC with a logic-level clock input signal to externally set the converter's operating frequency. DC-DC conversion cycles initiate on the rising edge of the input clock signal. Note that when driving SYNC with an external signal, RFREQ must still be connected to FREQ.
POL VL
2
3
FREQ
3 4 5 -- -- 6 7 8 9 10
4 5 6 7,9 8 10,11 12 13 14 15
COMP REF FB N.C. SHDN GND PGND CS EXT IN
--
16
SYNC
8
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High-Efficiency, Current-Mode, Inverting PWM Controller
Typical Application Circuit
VIN +3V to +5.5V 3 x 22F 10V
MAX1846/MAX1847
22k FDS6375 CMSH5-40 0.47F 2 VL 8 SHDN IN 14 13 Sanyo 16TPB47M 7, 9 0.02 1W R1 95.3k 1% 10H DO5022P-103 47F 16V 47F 16V 15
VOUT -12V AT 400mA
EXT CS
16
SYNC
MAX1847
4 10k 0.22F 3 5 150k FREQ REF POL 1 GND COMP
N.C.
PGND
12
FB
6
R2 10.0k 1% 1200pF
10, 11
0.1F
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
Functional Diagram
IN
EXT SHDN MAX1847 ONLY STARTUP CIRCUITRY PGND EXT DRIVER VL REGULATOR VL
UNDERVOLTAGE LOCK OUT POL
CONTROL CIRCUITRY
MAX1846 MAX1847
SYNC MAX1847 ONLY FREQ
OSCILLATOR
ERROR COMPARATOR COMP
FB GM ERROR AMPLIFIER SOFT-START SLOPE COMP REF REFERENCE CURRENTSENSE AMPLIFIER
CS
PGND
GND
10
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High-Efficiency, Current-Mode, Inverting PWM Controller
Detailed Description
The MAX1846/MAX1847 current-mode PWM controller use an inverting topology that is ideal for generating output voltages from -2V to -200V. Features include shutdown, adjustable internal operating frequency or synchronization to an external clock, soft-start, adjustable current limit, and a wide (+3V to +16.5V) input range. 2) when exiting shutdown with power already applied, and 3) when exiting undervoltage lockout.
MAX1846/MAX1847
Shutdown (MAX1847 only)
The MAX1847 shuts down to reduce the supply current to 10A when SHDN is low. In this mode, the internal reference, error amplifier, comparators, and biasing circuitry turn off. The EXT output becomes high impedance and the external pullup resistor connected to EXT pulls VEXT to VIN, turning off the P-channel MOSFET. When in shutdown mode, the converter's output goes to 0.
PWM Controller
The architecture of the MAX1846/MAX1847 currentmode PWM controller is a Bi-CMOS multi-input system that simultaneously processes the output-error signal, the current-sense signal, and a slope-compensation ramp (Functional Diagram). Slope compensation prevents subharmonic oscillation, a potential result in current-mode regulators operating at greater than 50% duty cycle. The controller uses fixed-frequency, current-mode operation where the duty ratio is set by the input-to-output voltage ratio. The current-mode feedback loop regulates peak inductor current as a function of the output error signal.
Frequency Synchronization (MAX1847 only)
The MAX1847 is capable of synchronizing its switching frequency with an external clock source. Drive SYNC with a logic-level clock input signal to synchronize the MAX1847. A switching cycle starts on the rising edge of the signal applied to SYNC. Note that the frequency of the signal applied to SYNC must be higher than the default frequency set by RFREQ. This is required so that the internal clock does not start a switching cycle prematurely. If SYNC is inactive for an entire clock cycle of the internal oscillator, the internal oscillator takes over the switching operation. Choose RFREQ such that fOSC = 0.9 fSYNC.
Internal Regulator
The MAX1846/MAX1847 incorporate an internal lowdropout regulator (LDO). This LDO has a 4.25V output and powers all MAX1846/MAX1847 functions (excluding EXT) for the primary purpose of stabilizing the performance of the IC over a wide input voltage range (+3V to +16.5V). The input to this regulator is connected to IN, and the dropout voltage is typically 100mV, so that when VIN is less than 4.35V, VL is typically VIN minus 100mV. When the LDO is in dropout, the MAX1846/MAX1847 still operate with VIN as low as 3V. For best performance, it is recommended to connect VL to IN when the input supply is less than 4.5V.
EXT Polarity (MAX1847 only)
The MAX1847 features an option to utilize an N-channel MOSFET configuration, rather than the typical P-channel MOSFET configuration (Figure 1). In order to drive the different polarities of these MOSFETs, the MAX1847 is capable of reversing the phase of EXT by 180 degrees. When driving a P-channel MOSFET, connect POL to GND. When driving an N-Channel MOSFET, connect POL to VL. These POL connections ensure the proper polarity for EXT. For design guidance in regard to this application, refer to the MAX1856 data sheet.
Undervoltage Lockout
The MAX1846/MAX1847 have an undervoltage lockout circuit that monitors the voltage at VL. If VL falls below the UVLO threshold (2.8V typ), the control logic turns the P-channel FET off (EXT high impedance). The rest of the IC circuitry is still powered and operating. When VL increases to 60mV above the UVLO threshold, the IC resumes operation from a start up condition (soft-start).
Design Procedure
Initial Specifications
In order to start the design procedure, a few parameters must be identified: the minimum input voltage expected (V IN(MIN) ), the maximum input voltage expected (VIN(MAX)), the desired output voltage (VOUT), and the expected maximum load current (ILOAD). Calculate the Equivalent Load Resistance This is a simple calculation used to shorten the verification equations: RLOAD = VOUT / ILOAD
Soft-Start
The MAX1846/MAX1847 feature a "digital" soft-start that is preset and requires no external capacitor. Upon startup, the FB threshold decrements from the reference voltage to 0 in 64 steps over 1024 cycles of fOSC or fSYNC. See the Typical Operating Characteristics for a scope picture of the soft-start operation. Soft-start is implemented: 1) when power is first applied to the IC,
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11
High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
VIN +12V 12F VP1-0190 25V 12.2H
1:4
CMR1U-02 0.47F 8 16 1 2 VL POL SHDN SYNC CS MAX1847 4 0.033F 270k 3 5 COMP FREQ REF GND 10, 11 1800pF FB 6 10.0k 1% N.C. PGND 15 IN 14 EXT 13 7, 9 12 0.05 0.5W 100pF 100V 383k 1% IRLL2705 470
VOUT -48V AT 100mA 12F 100V
150k
0.1F
Figure 1. Using an N-Channel MOSFET (MAX1847 only)
Calculate the Duty Cycle The duty cycle is the ratio of the on-time of the MOSFET switch to the oscillator period. This is determined by the ratio of the input voltage to the output voltage. Since the input voltage typically has a range of operation, a minimum (DMIN) and maximum (DMAX) duty cycle is calculated by: DMIN =
- VOUT + VD VIN(MAX) - VSW - VLIM - VOUT + VD - VOUT + VD VIN(MIN) - VSW - VLIM - VOUT + VD
1.25V and the regulation voltage for FB is nominally 0. The load presented to the reference by the feedback resistors must be less than 500A. This is to guarantee that VREF is in regulation (see Electrical Characteristics Table). Conversely, the current through the feedback resistors must be large enough so that the leakage current of the FB input (50nA) is insignificant. Therefore, select R2 so that IR2 is between 50A and 250A. IR2 = VREF / R2 where VREF = 1.25V. A typical value for R2 is 10k. Once R2 is selected, calculate R1 with the following equation: R1 = R2 x (-VOUT / VREF)
DMAX =
where VD is the forward drop across the output diode, VSW is the drop across the external FET when on, and V LIM is the current-limit threshold. To begin with, assume VD = 0.5V for a Schottky diode, VSW = 100mV, and VLIM = 100mV. Remember that VOUT is negative when using this formula.
Setting the Operating Frequency
The MAX1846/MAX1847 are capable of operating at switching frequencies from 100kHz to 500kHz. Choice of operating frequency depends on a number of factors: 1) Noise considerations may dictate setting (or synchronizing) f OSC above or below a certain frequency or band of frequencies, particularly in RF applications.
Setting the Output Voltage
The output voltage is set using two external resistors to form a resistive-divider to FB between the output and REF (refer to R1 and R2 in Figure 1). VREF is nominally
12
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High-Efficiency, Current-Mode, Inverting PWM Controller
2) 3) Higher frequencies allow the use of smaller value (hence smaller size) inductors and capacitors. Higher frequencies consume more operating power both to operate the IC and to charge and discharge the gate of the external FET. This tends to reduce the efficiency at light loads. Higher frequencies may exhibit lower overall efficiency due to more transition losses in the FET; however, this shortcoming can often be nullified by trading some of the inductor and capacitor size benefits for lower-resistance components. High-duty-cycle applications may require lower frequencies to accommodate the controller minimum off-time of 0.4s. Calculate the maximum oscillator frequency with the following formula: fOSC(MAX) = VIN(MIN) - VSW - VLIM tions, most circuits are more efficient and economical operating in continuous mode, which refers to continuous current in the inductor. In continuous mode there is a trade-off between efficiency and transient response. Higher inductance means lower inductor ripple current, lower peak current, lower switching losses, and, therefore, higher efficiency. Lower inductance means higher inductor ripple current and faster transient response. A reasonable compromise is to choose the ratio of inductor ripple current to average continuous current at minimum duty cycle to be 0.4. Calculate the inductor ripple with the following formula: IRIPPLE = 0.4 x ILOAD(MAX) x VIN(MAX) - VSW - VLIM - VOUT + VD
MAX1846/MAX1847
4)
5)
(VIN(MAX) - VSW - VLIM )
(
)
VIN(MIN) - VSW - VLIM - VOUT + VD 1 x t OFF(MIN)
Then calculate an inductance value: L = (VIN(MAX) / IRIPPLE) x (DMIN / fOSC) Choose the closest standard value. Once again, remember that VOUT is negative when using this formula.
Remember that VOUT is negative when using this formula. The oscillator frequency is set by a resistor, RFREQ, connected from FREQ to GND. The relationship between fOSC (in Hz) and RFREQ (in ) is slightly nonlinear, as illustrated in the Typical Operating Characteristics. Choose the resistor value from the graph and check the oscillator frequency using the following formula:
1
Determining Peak Inductor Current
The peak inductor current required for a particular output is: ILPEAK = ILDC + (ILPP / 2) where ILDC is the average DC input current and ILPP is the inductor peak-to-peak ripple current. The ILDC and ILPP terms are determined as follows: ILDC = ILOAD x ( - VOUT + VD ) VIN(MIN) - VSW - VLIM
fOSC =
2 -19 5.21 x 10 -7 + 1.92 x 10 -11 x R x (RFREQ ) FREQ - 4.86 x 10
(
)(
)
(
)
ILPP = External Synchronization (MAX1847 only) The SYNC input provides external-clock synchronization (if desired). When SYNC is driven with an external clock, the frequency of the clock directly sets the MAX1847's switching frequency. A rising clock edge on SYNC is interpreted as a synchronization input. If the sync signal is lost, the internal oscillator takes over at the end of the last cycle, and the frequency is returned to the rate set by RFREQ. Choose RFREQ such that fOSC = 0.9 x fSYNC.
(VIN(MIN) - VSW - VLIM ) x (- VOUT + VD )
L x fOSC x ( - VOUT + VD )
Choosing Inductance Value
The inductance value determines the operation of the current-mode regulator. Except for low-current applica-
where L is the selected inductance value. The saturation rating of the selected inductor should meet or exceed the calculated value for ILPEAK, although most coil types can be operated up to 20% over their saturation rating without difficulty. In addition to the saturation criteria, the inductor should have as low a series resistance as possible. For continuous inductor current, the power loss in the inductor resistance (PLR) is approximated by: PLR ~ (ILOAD x VOUT / VIN)2 x RL where RL is the inductor series resistance.
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13
High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
Once the peak inductor current is calculated, the current sense resistor, RCS, is determined by: RCS = 85mV / ILPEAK For high peak inductor currents (>1A), Kelvin-sensing connections should be used to connect CS and PGND to R CS . Connect PGND and GND together at the ground side of RCS. A lowpass filter between RCS and CS may be required to prevent switching noise from tripping the current-sense comparator at heavy loads. Connect a 100 resistor between CS and the high side of RCS, and connect a 1000pF capacitor between CS and GND. associated with charging the gate. In addition, this parameter helps predict the current needed to drive the gate at the selected operating frequency. The power MOSFET in an inverting converter must have a high enough voltage rating to handle the input voltage plus the magnitude of the output voltage and any spikes induced by leakage inductance. Choose RDS(ON)(MAX) specified at VGS < VIN(MIN) to be one to two times RCS. Verify that VIN(MAX) < VGS(MAX) and VDS(MAX) > VIN(MAX) - VOUT + VD. Choose the riseand fall-times (tR, tF) to be less than 50ns.
Output Capacitor Selection
The output capacitor (COUT) does all the filtering in an inverting converter. The output ripple is created by the variations in the charge stored in the output capacitor with each pulse and the voltage drop across the capacitor's equivalent series resistance (ESR) caused by the current into and out of the capacitor. There are two properties of the output capacitor that affect ripple voltage: the capacitance value, and the capacitor's ESR. The output ripple due to the output capacitor's value is given by: VRIPPLE-C = (ILOAD DMAX TOSC ) / COUT The output ripple due to the output capacitor's ESR is given by: VRIPPLE-R = ILPP RESR These two ripple voltages are additive and the total output ripple is: VRIPPLE-T = VRIPPLE-C + VRIPPLE-R The ESR-induced ripple usually dominates this last equation, so typically output capacitor selection is based mostly upon the capacitor's ESR, voltage rating, and ripple current rating. Use the following formula to determine the maximum ESR for a desired output ripple voltage (VRIPPLE-D): RESR = VRIPPLE-D / ILPP Select a capacitor with ESR rating less than RESR. The value of this capacitor is highly dependent on dielectric type, package size, and voltage rating. In general, when choosing a capacitor, it is recommended to use low-ESR capacitor types such as ceramic, organic, or tantalum capacitors. Ensure that the selected capacitor has sufficient margin to safely handle the maximum ripple current (ILPP) and the maximum output voltage.
Checking Slope-Compensation Stability
In a current-mode regulator, the cycle-by-cycle stability is dependent on slope compensation to prevent subharmonic oscillation at duty cycles greater than 50%. For the MAX1846/MAX1847, the internal slope compensation is optimized for a minimum inductor value (LMIN) with respect to duty cycle. For duty cycles greater then 50%, check stability by calculating LMIN using the following equation: LMIN = VIN(MIN) x RCS / MS
x (2 x DMAX - 1) / (1 - DMAX )
[
[(
)
]
]
where VIN(MIN) is the minimum expected input voltage, Ms is the Slope Compensation Ramp (41 mV/s) and DMAX is the maximum expected duty cycle. If LMIN is larger than L, increase the value of L to the next standard value that is larger than L MIN to ensure slope compensation stability.
Power MOSFET Selection
The MAX1846/MAX1847 drive a wide variety of P-channel power MOSFETs (PFETs). The best performance, especially with input voltages below 5V, is achieved with low-threshold PFETs that specify on-resistance with a gate-to-source voltage (VGS) of 2.7V or less. When selecting a PFET, key parameters include: 1) Total gate charge (QG) 2) Reverse transfer capacitance (CRSS) 3) On-resistance (RDS(ON)) 4) Maximum drain-to-source voltage (VDS(MAX)) 5) Minimum threshold voltage (VTH(MIN)) At high switching rates, dynamic characteristics (parameters 1 and 2 above) that predict switching losses may have more impact on efficiency than R DS(ON), which predicts DC losses. QG includes all capacitance
14
Choosing Compensation Components
The MAX1846/MAX1847 are externally loop-compensated devices. This provides flexibility in designs to accommodate a variety of applications. Proper com-
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High-Efficiency, Current-Mode, Inverting PWM Controller
pensation of the control loop is important to prevent excessive output ripple and poor efficiency caused by instability. The goal of compensation is to cancel unwanted poles and zeros in the DC-DC converter's transfer function created by the power-switching and filter elements. More precisely, the objective of compensation is to ensure stability by ensuring that the DCDC converter's phase shift is less than 180 by a safe margin, at the frequency where the loop gain falls below unity. One method for ensuring adequate phase margin is to introduce corresponding zeros and poles in the feedback network to approximate a single-pole response with a -20dB/decade slope all the way to unity-gain crossover. Calculating Poles and Zeros The MAX1846/MAX1847 current-mode architecture takes the double pole caused by the inductor and output capacitor and shifts one of these poles to a much higher frequency. This makes loop compensation easier. To compensate these devices, we must know the center frequencies of the right-half plane zero (zRHP) and the higher frequency pole (pOUT2). Calculate the zRHP frequency with the following formula: 2 - (1 - D MAX ) x VIN(MIN) - VOUT x RLOAD ZRHP = 2 x VOUT x L) (
2 G x R x 1 - D O( MAX ) x VIN(MIN) - VOUT M x R LOAD ADC = R x VIN(MIN) + TOSC (1 - DMAX ) CS Bx VIN(MIN) x RLOAD x RCS + MS1 2L
(
)
MAX1846/MAX1847
(
)
where: B is the feedback divider attenuation factor = (-VOUT / VREF), G M is the error amplifier transconductance = 400 A/V, RO is the error amplifier output resistance = 3 M, M S1 is the slope compensation factor = [(1.636A / s) RCS], RCS is the selected current sense resistor, L is the selected inductance value If zRHP is at a lower frequency than pOUT2, the required dominant pole frequency is given by: pDOM = zRHP / ADC Otherwise the required dominant pole frequency is: pDOM = pOUT2 / ADC Determining the Compensation Component Values Using p DOM, calculate the compensation capacitor required: CCOMP = 1 / (2 RO pDOM) Select the next largest standard value of capacitor and then calculate the compensation resistor required to cancel out the output-capacitor-induced pole (pOUT1) determined previously. A zero is needed to cancel the output-induced pole and the frequency of this zero must equal pOUT1. Therefore: zCOMP = pOUT1 RCOMP = RLOAD COUT / CCOMP Choose the nearest lower standard value of the resistor. Now check the final values selected for the compensation components: pCOMP = 1 / [2 CCOMP x (RO + RCOMP)] In order for pCOMP to compensate the loop, the openloop gain must reach unity at a lower frequency than the right-half-plane zero or the second output pole, whichever is lower in frequency. If the second output pole and the right-half-plane zero are close together in frequency, the higher resulting phase shift at unity gain
15
(
)
The calculations for pOUT2 are very complex. For most applications where VOUT does not exceed -48V (in a negative sense), the pOUT2 will not be lower than 1/8th of the oscillator frequency and is generally at a higher frequency than zRHP. Therefore: pOUT2 0.125 fOSC A pole is created by the output capacitor and the load resistance. This pole must also be compensated and its center frequency is given by the formula: pOUT1 = 1 / (2 RLOAD COUT) Finally, there is a zero introduced by the ESR of the output capacitor. This zero is determined from the following equation: zESR = 1 / (2 COUT RESR) Calculating the Required Pole Frequency To ensure stability of the MAX1846/MAX1847, the introduced pole (PDOM) by the compensation network must roll-off the error amplifier gain to 1 before z RHP or POUT2 occurs. First calculate the DC open-loop gain to determine the frequency of the pole to introduce.
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
may require a larger compensation capacitor than calculated. It might take more than a couple of iterations to obtain a suitable combination. Finally, the zero introduced by the output capacitor's ESR must be compensated. This is accomplished by placing a capacitor between REF and FB creating a pole directly in the feedback loop. Calculate the value of this capacitor using the frequency of zESR and the selected feedback resistor values with the formula: R + R2 CFB = RESR x COUT x 1 R1 x R2 exceed the potential difference between VOUT and the input voltage plus the leakage inductance spikes. For high output voltages (-50V or more), Schottky diodes may not be practical because of this voltage requirement. In these cases, use an ultrafast recovery diode with adequate reverse-breakdown voltage.
Input Filter Capacitor
The input capacitor (CIN) in inverting converter designs reduces the current peaks drawn from the input supply and reduces noise injection. The source impedance of the input supply largely determines the value of CIN. High source impedance requires high input capacitance, particularly as the input voltage falls. Since inverting converters act as "constant-power" loads to their input supply, input current rises as the input voltage falls. Consequently, in low-input-voltage designs, increasing C IN and/or lowering its ESR can add as much as 5% to the conversion efficiency.
Applications Information
Maximum Output Power
The maximum output power that the MAX1846/MAX1847 can provide depends on the maximum input power available and the circuit's efficiency: POUT(MAX) = Efficiency PIN(MAX) Furthermore, the efficiency and input power are both functions of component selection. Efficiency losses can be divided into three categories: 1) resistive losses across the inductor, MOSFET on-resistance, currentsense resistor, and the ESR of the input and output capacitors; 2) switching losses due to the MOSFET's transition region, and charging the MOSFET's gate capacitance; and 3) inductor core losses. Typically 80% efficiency can be assumed for initial calculations. The required input power depends on the inductor current limit, input voltage, output voltage, output current, inductor value, and the switching frequency. The maximum output power is approximated by the following formula: PMAX = [VIN - (VLIM + ILIM x RDS(ON))] x ILIM x [1 - (LIR / 2)] x [(-VOUT + VD) / (VIN - VSW - VLIM - VOUT + VD)] where I LIM is the peak current limit and LIR is the inductor current-ripple ratio and is calculated by: LIR = ILPP / ILDC Again, remember that V OUT for the MAX1846/ MAX1847 is negative.
Bypass Capacitor
In addition to CIN and COUT, other ceramic bypass capacitors are required with the MAX1846/MAX1847. Bypass REF to GND with a 0.1F or larger capacitor. Bypass VL to GND with a 0.47F or larger capacitor. All bypass capacitors should be located as close to their respective pins as possible.
PC Board Layout Guidelines
Good PC board layout and routing are required in highfrequency-switching power supplies to achieve good regulation, high efficiency, and stability. It is strongly recommended that the evaluation kit PC board layouts be followed as closely as possible. Place power components as close together as possible, keeping their traces short, direct, and wide. Avoid interconnecting the ground pins of the power components using vias through an internal ground plane. Instead, keep the power components close together and route them in a "star" ground configuration using component-side copper, then connect the star ground to internal ground using multiple vias.
Main Application Circuits
The MAX1846/MAX1847 are extremely versatile devices. Figure 2 shows a generic schematic of the MAX1846. Table 1 lists component values for several typical applications. These component values also apply to the MAX1847. The first two applications are featured in the MAX1846/MAX1847 EV Kit.
Diode Selection
The MAX1846/MAX1847's high-switching frequency demands a high-speed rectifier. Schottky diodes are recommended for most applications because of their fast recovery time and low forward voltage. Ensure that the diode's average current rating exceeds the peak inductor current by using the diode manufacturer's data. Additionally, the diode's reverse breakdown voltage must
16
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High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
VIN
APPLICATION B ONLY
CIN
22k P D1
0.47F
1 VL
10 IN EXT CS MAX1846 RCS R1 9 8 L1
VOUT COUT
3 CCOMP RCOMP RFREQ 2 4
COMP FREQ REF
PGND
7
FB GND 6
5 R2 CFB
0.1F
NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS.
Figure 2. MAX1846 Main Application Circuit
Table 1. Component List for Main Application Circuits
CIRCUIT ID Input (V) Output (V) Output (A) CCOMP (F) CIN (F) COUT (F) CFB (pF) R1 (k) (1%) R2 (k) (1%) RCOMP (k) RCS () RFREQ (k) D1 L1 (H) P1 A 12 -5 2 0.047 3 x 10 2 x 100 390 40.2 10 8.2 0.02 150 CMSH5-40 10 FDS6685 B 3 to 5.5 -12 0.4 0.22 3 x 22 2 x 47 1200 95.3 10 10 0.02 150 CMSH5-40 10 FDS6375 C 12 -48 0.1 0.068 10 47 1800 383 10 150 0.05 150 CMR1U-02 47 IRFR5410 D 12 -72 0.1 0.1 10 33 1800 576 10 1800 0.05 150 CMR1U-02 82 IRFR5410
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17
High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
Component Suppliers
SUPPLIER AVX Central Semiconductor Coilcraft Dale Fairchild International Rectifier IRC Kemet On Semiconductor Panasonic Sanyo Siliconix Sprague Sumida Vitramon COMPONENT Capacitors Diodes Inductors Resistors MOSFETs MOSFETs Resistors Capacitors MOSFETs, Diodes Capacitors, Resistors Capacitors MOSFETs Capacitors Inductors Resistors PHONE 803-946-0690 516-435-1110 847-639-6400 402-564-3131 408-721-2181 310-322-3331 512-992-7900 864-963-6300 602-303-5454 201-348-7522 619-661-6835 408-988-8000 603-224-1961 847-956-0666 203-268-6261 WEBSITE www.avxcorp.com www.centralsemi.com www.coilcraft.com www.vishay.com/brands/dale/main.html www.fairchildsemi.com www.irf.com www.irctt.com www.kemet.com www.onsemi.com www.panasonic.com www.secc.co.jp www.siliconix.com www.vishay.com/brands/sprague/main.html www.remtechcorp.com www.vishay.com/brands/vitramon/main.html
Note: Please indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers.
Pin Configurations
TOP VIEW
VL 1 FREQ COMP REF FB 2 3 4 5 10 IN 9 EXT CS POL 1 VL 2 FREQ 3 16 SYNC 15 IN 14 EXT
Chip Information
TRANSISTOR COUNT: 2441 PROCESS TECHNOLOGY: BiCMOS
MAX1846
8 7 6
PGND COMP 4 GND REF 5 FB 6 N.C. 7 SHDN 8
MAX1847
13 CS 12 PGND 11 GND 10 GND 9 N.C.
10-PIN MAX
16-PIN QSOP
18
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High-Efficiency, Current-Mode, Inverting PWM Controller
Package Information
10LUMAX.EPS
MAX1846/MAX1847
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19
High-Efficiency, Current-Mode, Inverting PWM Controller MAX1846/MAX1847
Package Information (continued)
QSOP.EPS
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2001 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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